GSM Technique

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DIPLOMARBEIT

Interference Reductionin a GSM Handset

ausgefhrt zum Zwecke der Erlangung des akademischen Gradeseines Diplom-Ingenieurs unter Zusammenarbeit von

INSTITUTFR NACHRICHTENTECHNIKUND HOCHFREQUENZTECHNK

TECHNISCE UNIVERSITT WIENAUSTRIA

Center For PersonKommunikationAalborg University, Denmark

eingereicht an der Technischen Universitt WienFakultt fr Elektrotechnik

von

Thomas Baumgartner

Gr. Kadolz 94A-2062 Seefeld-Kadolz

Matr.Nr.: 9426012e-mail: [email protected]

Wien, im August 1999


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Supervisors:M.SC.E.E. Gert F. Pedersen, CPK-Aalborg University

M.SC.E.E. Mikael B. Knudsen, Bosch Telekom Danmark A/So.Univ.Prof. Dipl.-Ing. Dr. Ernst Bonek, INTHFT/TU-Wien

Dipl.-Ing. Thomas Neubauer, INTHFT/TU-Wien


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Zusammenfassung

In den letzten Jahren erregten adaptive Antennen fr Basisstationenen regesInteresse. Verschiedene Hersteller, Netzwerkbetreiber und Universitten fhrtenFeldversuche durch um mehr Informationen ber die Leistungsfhigkeit dieser

Systeme zu erhalten. Feldversuche in GSM Netzwerken haben gezeigt, da fr denDownlink sowohl in Makro- als auch Mikrozellen eine bemerkenswerteVerbesserung fr C/N (Carrier zu Noise Verhltnis) und C/I (Carrier zuInterference Verhltnis). Fr den Downlink wurden signifikante Verbesserungen inMakrozellenumgebungen festgestellt. In Mikrozellenumgebungen wurde jedochnur ein kleiner Gewinn fr C/N und so gut wie keine C/I Verbesserung gemessen.

Diese Diplomarbeit ist darauf ausgerichtet einen passenden Algorithmus fr einAntennensystem mit geringer Komplexitt, das in einem Mobiltelefon eingesetztwerden kann, zu finden und zu simulieren. Dieses System soll das C/I imDownlink in einer Mikrozellenumgebung mit sich langsam bewegenden Bentzernverbessern.

Es gibt verschiedene Methoden ein Antennensystem fr diesen Zweck zuimplementieren. Fr diese Arbeit wurde ein Antennensystem mit 2Antennenelementen, wo die Amplitude und die Phase eines Antennenzweiges mitHilfe eines variablen Verstrkers und eines Phasenschiebers adaptiert wird, bevordie Antennensignale zusammengefhrt werden. Die Komplexitt dieses System istgering genug um es in einem Mobiltelefon einzusetzen, da lediglich einezustzliche Antenne, ein variabler Verstrker, ein Phasenschieber und einSummierglied bentigt werden.

Unter Bercksichtigung der Resultate von Hagerman [Hag95] und der groenKohrenzbandbreite der gemessenen Kanaldaten wurde die Simulation des

gefundenen Algorithmus auf einen Gleichkanalstrer und einen Kanal mit flachemFading beschrnkt.


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Abstract

Adaptive antennas for base stations have obtained great interest over the past fewyears and at present. Several manufacturers, operators and universities have and areperforming field test to get more detailed information about the performance of

such systems. GSM field tests have shown a significant improvement of the uplinkperformance for both C/N (Carrier to Noise ratio) and C/I (Carrier to Interferenceratio) in both macro cell and micro cell applications. For the downlink significantimprovements are observed for macro cell environments, but for micro cellenvironments only small gains for C/N and almost no C/I improvements areobserved.

This project is concentrated about finding and simulating a suitable combiningalgorithm for a low complexity antenna system for a GSM mobile handset, whichcan improve the downlink C/I performance in a micro cell environment with slowmoving users.

There are several methods for implementing an antenna system to improve thedownlink C/I performance. For this project an antenna system consisting of 2antenna elements, where the amplitude and phase of the received signals from oneof the antennas are altered prior to a combining by use of a variable gain block anda phase shifter right after the antenna element. The complexity of this antennasystem is low enough to be suitable for a mobile handset, hence it only requires anextra antenna element, a phase shifter and a combiner compared to a standardmobile handset without any antenna system.

Taking the results of Hagerman [Hag95] and the huge coherence bandwidth of themeasured channel data into account the simulations are limited to 1 co-channelinterferer and a flat fading channel.


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Preface

This thesis presents the research work done at the Department of CommunicationsTechnology at Aalborg University. It is part of an ERASMUS student exchangeprogram with my home university, Technische Universitt Wien, Austria.

The thesis is divided into the following parts:

A description of the GSM system. A brief description of radio wave propagation. A description of different diversity schemes and combining techniques. An explanation of how the necessary parameters for the chosen combining

algorithm can be estimated from the received signal. A description of the receiver structure in GSM and the necessary changes in

order to implement the proposed combining method. A description of the simulated algorithm. A presentation of the simulation results. A suggestion for a real time test configuration.

Appendices are placed in the last part, their purpose is to give supplementaryinformation when reading the report.


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Acknowledgement

I want to thank my supervisors, Gert Frlund Pedersen and Mikael BergholzKnudsen for the fruitful discussions and their invaluable expert advice. Thanks toThomas Neubauer who gave me a lot of useful advice for my stay in Aalborg.

Thanks are also extended to Nina Nielsen at CPK.

I am pleased to acknowledge the financial support of the SOKRATES/ERASMUSexchange program and of the Siegfried Ludwig-Fonds fr universitreEinrichtungen in Niedersterreich, who made my stay in Aalborg possible.

Special thanks to Dieter Schafhuber for solving bureaucratic problems in Viennaduring my stay in Denmark and to Martin Pillwatsch for his friendship. Thanks toSabine for her love understanding and patience during my long absence fromhome.

Aalborg, June 1999

Thomas Baumgartner


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Table of Contents

CHAPTER 1 GSM SYSTEM....................................................................................... 1

1.1 Historical Note .......................................................................................................................................1

1.2 Cell Structure.........................................................................................................................................2

1.3 GSM Network ........................................................................................................................................2

1.4 The Radio Interface...............................................................................................................................31.4.1 Bursts and Synchronisation .............................................................................................................41.4.2 Logical channels ..............................................................................................................................51.4.3 Frequency Hopping..........................................................................................................................61.4.4 Discontinuous Transmission............................................................................................................81.4.5 Power Control..................................................................................................................................8

1.5 Frame Structure.....................................................................................................................................8

1.6 Channel Coding .....................................................................................................................................91.6.1 Coding .............................................................................................................................................9

1.6.2 Interleaving......................................................................................................................................91.6.3 Modulation.....................................................................................................................................10

CHAPTER 2 RADIO WAVE PROPAGATION.......................................................... 11

2.1 The Physics of Radio Wave Propagation...........................................................................................112.1.1 Reflection.......................................................................................................................................112.1.2 Diffraction......................................................................................................................................132.1.3 Scattering.......................................................................................................................................132.1.4 Free Space Propagation .................................................................................................................14

2.2 Multipath Propagation........................................................................................................................15

2.2.1 Slow Fading...................................................................................................................................152.2.2 Fast Fading.....................................................................................................................................162.2.3 Doppler Shift .................................................................................................................................202.2.4 Delay Spread..................................................................................................................................22

CHAPTER 3 DIVERSITY AND COMBINING METHODS........................................ 25

3.1 Different Diversity Schemes................................................................................................................253.1.1 Space Diversity..............................................................................................................................253.1.2 Polarisation Diversity ....................................................................................................................263.1.3 Pattern Diversity ............................................................................................................................273.1.4 Frequency Diversity.......................................................................................................................27

3.1.5 Time Diversity...............................................................................................................................28

3.2 Combining Techniques........................................................................................................................283.2.1 Switched Combining......................................................................................................................28


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Table of Contents viii

3.2.2 Selection Combining......................................................................................................................293.2.3 Maximum Ratio Combining ..........................................................................................................293.2.4 Equal Gain Combining ..................................................................................................................293.2.5 Optimum Combining.....................................................................................................................29

3.3 Model For Optimum Combining........................................................................................................30

3.3.1 Algorithm A...................................................................................................................................333.3.2 Algorithm B...................................................................................................................................33

CHAPTER 4 SIGNAL ESTIMATION........................................................................ 35

4.1 Signal Composition..............................................................................................................................35

4.2 Propagation Vector of Wanted Signal ...............................................................................................38

4.3 Propagation Vector of Interfering Signal ..........................................................................................40

4.4 Detection of Interferer Position ..........................................................................................................42

4.5 Problematic Positions of the Bursts....................................................................................................434.5.1 Only a part of the interfering training sequence overlaps with the wanted burst...........................444.5.2 Change of interferer power during wanted time slot .....................................................................47

4.6 Noise Power..........................................................................................................................................47

CHAPTER 5 RECEIVER STRUCTURE................................................................... 49

5.1 Classical GSM Receiver ......................................................................................................................495.1.1 RF Stage ........................................................................................................................................505.1.2 IF Stage..........................................................................................................................................50

5.1.3 Quadrature Stage ...........................................................................................................................505.1.4 Digitalisation stage ........................................................................................................................505.1.5 Detection stage...............................................................................................................................505.1.6 Decoding Stage..............................................................................................................................52

5.2 Necessary Changes in the Receiver ....................................................................................................52

CHAPTER 6 TRACKING ALGORITHM................................................................... 55

6.1 Introduction .........................................................................................................................................55

6.2 Block Diagram of Algorithm...............................................................................................................57

CHAPTER 7 SIMULATION RESULTS .................................................................... 61

7.1 Optimal Case........................................................................................................................................61

7.2 Different Antenna Powers...................................................................................................................64

7.3 Effect of Unsynchronised Network.....................................................................................................667.3.1 Overlapping Training Sequences...................................................................................................667.3.2 Overlapping Guard Period.............................................................................................................677.3.3 Gain Over Position of Interfering Training Sequence ...................................................................68

CHAPTER 8 REAL TIME TEST............................................................................... 71

8.1 Test Setup.............................................................................................................................................71


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Table of Contents ix

8.2 Test Procedure .....................................................................................................................................72

CONCLUSION.......................................................................................................... 73

ABBREVIATIONS.................................................................................................... 75

BIBLIOGRAPHY ...................................................................................................... 77

APPENDIX A GENERATION OF A GSM BURST..................................................A-1

APPENDIX B DATA USED FOR THE SIMULATIONS...........................................B-1

B.1 Measurements ....................................................................................................................................B-1

B.2 Data Processing..................................................................................................................................B-3B.2.1 Data Used for Determining the Weight Set .................................................................................B-3B.2.2 Data Used for All Other Simulations...........................................................................................B-3

APPENDIX C TRAINING SEQUENCES IN GSM ...................................................C-1

APPENDIX D WEIGHT SET ...................................................................................D-1

APPENDIX E ESTIMATION OF INPUT SIGNAL....................................................E-1

E.1 Estimation Necessary for Algorithm A............................................................................................E-2

E.2 Estimation Necessary for Algorithm B ............................................................................................E-5

APPENDIX F COMPARISON ALGORITHM A AND B........................................... F-1

F.1 Ideal Conditions.................................................................................................................................F-1

F.2 Real Conditions.................................................................................................................................. F-3


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1

Chapter 1

GSM System

This chapter gives a brief overview of the GSM system. The GSM family contains GSM900,DCS1800 and PCS1900 whose main difference is the used frequency band.The first part of this chapter (Section 1.1 to 1.3) give an overall description of the GSM system and thefollowing sections deal with more detailed information about the burst structure and the channelcoding.

1.1 Historical NoteIn the early 1980s there were several different, to each other incompatible analog cellular phonesystems in Europe. International roaming was nearly impossible because each country developed itsown system. Therefore the costs for the equipment were very high due to the small market for eachsystem type.This was recognised very early and in 1982 the Conference of European Posts and Telegraphs (CEPT)formed the Groupe Spcial Mobile (GSM). The task for this group was to develop a pan-Europeanpublic land mobile system. The system had to meet certain criteria [Ste92]:

spectrum efficiency

subjective voice quality low mobile cost hand-portable feasibility low base station cost ability to support new services co-existence with current systems

In 1989 the responsibility for GSM was transferred to the European Telecommunications StandardInstitute (ETSI). Phase 1 of the GSM specification was published 1990 and the first commercial GSMnetwork started their services mid 91. Since that time point the number of GSM networks and theirusers increased rapidly. At the end of 1998 there were more than 320 GSM Networks (includingDCS1800 and PCS1900) in 129 countries. Figure 1-1 shows the number of the subscribers from 1992

to 1998.

0

20

40

60

80

100

120

140

160

Dec 92 Dec 93 Dec 94 Dec 95 Dec 96 Dec 97 Dec 98

millionsubscribers

Figure 1-1: The number of GSM subscribers from 1992 to 1998 [GSM99]


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1.2 Cell Structure 2

1.2 Cell StructureIn GSM the covered area is divided into cells. A base station transceiver (BTS) is placed in each cell.

A certain number of cells is grouped into cell clusters. The cells of one cluster share the available

carrier frequencies. The same carrier frequencies are reused in the cells of other clusters. Figure 1-2

shows a cluster with seven cells, numbered 1-7. Cells with the same number are called co-channel

cells because they use the same frequency sets.

Figure 1-2: The cellular structure

In rural areas with very low traffic is the size of the cells limited by the propagation loss, the

maximum transmitting power of the mobile stations (MS) and the propagation time. In urban areas

with high traffic volume it is tried to make the cells small in order to have a high number of traffic

channels per area. In this case the minimum cell size is given by the co-channel interference and thecosts which arise by having a lot of base stations with low transmitting power. Co-channel interference

means interference by a cell in a neighbouring cluster using the same frequencies.

It is optional whether the time base counters of different base stations are synchronised together. The

frequency accuracy of the frequency source of the base stations should be better than 0,05 ppm for RF

frequency generation and clocking the time base [GSM0510]

1.3 GSM NetworkThe GSM network can be divided into several functional parts whose functions and interfaces are

defined in the GSM specification. Figure 1-3 shows the general architecture of a GSM network.

The network can be divided into the operation subsystem, the radio subsystem and the network

subsystem. The radio subsystem consists of the mobile stations (MS) and one BTS for each cell. Theoperation subsystem contains the Operation and Maintenance Center (OMC) which handles

administrative tasks like billing and updating of the system. The network subsystem consists of the

Mobile Switching Center (MSC), the Visitor Location Register (VLR), the Home Location Register

(HLR), the Authentication Center (AC) and the Equipment Identifier Register (EIR).

The main part of the network system is the MSC which does the switching between mobile users and

between mobile users and users of other networks. The mobility management is although handled by

the MSC together with the MS, HLR and VLR.

In the HLR is subscriber specific data like subscribed services, mobile subscriber identity, directory

number, authentication code and the address of the VLR of all to the MSC subscribed users stored.

Data which is necessary for managing the MS is stored in the VLR for all MS which are in the area of

the MSC.

The EIR contains a list of all registered MSs by their International Mobile Equipment Identity (IMEI).With the help of this list lost, stolen or defect equipment can be recognised. In Dublin is the Central

Equipment Identity Register (CEIR) placed, where the identities of all mobile stations notified as


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1.4 The Radio Interface 3

approved, lost or stolen are stored in the "white" and "black" lists. From this location can all GSM

operators update their EIRs.

VLR HLR AuC EIR

MSC BSC BTS

BTS

OMC

PSTN/ISDN

datanetworks

Operation Subsystem Radio SubsystemOther Networks

physical connection logical connection

AuC Authentication CentreBSC Base Station ControllerBTS Base Transceiver StationEIR Equipment Identity RegisterHLR Home Location Register

MS Mobile StationMSC Mobile Switching CentreOMC Operation and Maintenance C.PSTN Public Switched Teleph. NetworkVLR Visitor Location Register

Network Subsystem

Figure 1-3: General architecture of a GSM network [Dav96]

1.4 The Radio InterfaceThe GSM system uses a combination of FDMA and TDMA with 8 time slots per carrier (see Figure

1-4). For the separation of up- and downlink Frequency Division Duplex (FDD) and Time Division

Duplex (TDD) are used. The time shift between the bursts of up- and downlink is 3 time slots. The

time shift was introduced in order to simplify the equipment because it is not necessary to send and

receive at the same time. The lower frequency band where attenuation in the channel is lower, was

chosen for the uplink. In the time between sending and receiving the MS monitors the power of other

Base Stations (BS).

Channel 4Channel 3

Channel 2Channel 1

Frequency

TDMA-Frame

Downlink

Uplink

TDD

(3 Slos)

FDD

(45MHz)

FDD Frequency Division Duplex

TDD Time Division Duplex

Channel 124

Figure 1-4: Schematic structure of the FDMA/TDMA radio interface [Dav96]


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1.4.1 Bursts and SynchronisationDuring the time slots the data is transmitted in packets with a given structure the so called bursts. As

shown in Figure 1-5 there are five types of bursts with a duration of

156,25 bits or 0,577 ms.

3 57 1 26 1 57 3 8,25

TB Data Bits SBTraining

SequenceSB Data Bits TB GP

Time Slot 156,25 bit

3 142 38,25

TB Fixed Bits TB GP

3 39 64 39 38,25

TB Data BitsTraining

SequenceData Bits TB GP

3 58 26 58 38,25

TB Mixed BitsTraining

SequenceMixed Bits TB GP

8 41 36 3 68,25

TB Training Sequence Data Bits TB GP

TB Tail BitsSB Stealing Bits

GP Guard Period

Normal Burst

Frequency Correction Channel Burst

Synchronisation Burst

Dummy Burst

Access Burst

Figure 1-5: The different bursts in GSM [Meh97]

Except the normal burst are all other bursts dedicated to a special function.

The Frequency Correction Channel Burst (FCCH) contains a plain sinus wave which is used to match

the carrier frequencies of the MS and the BS.

The Synchronisation Channel Burst (SCH) is used to achieve synchronisation in the time domain. The

64 bit long training sequence is known by the MS. The exact position of the bits can be recognised

through correlating the received training sequence with the stored version of this sequence. Then the

78 data bits are decoded which contain information about the actual frame number.

The normal burst is transmitted during a call-in-progress. The two blocks of 57 data bits contain

ciphered information. The 26 bit long training sequence is used for estimating the channel properties.

Another function of the training sequence is to distinct between signals from the wanted and the

interfering signal. In each cell cluster one of eight available training sequences is used. Therefore it is

possible to detect co-channel interferer by the different training sequence. The stealing bits which

guard the training sequence indicate if the information bits contain data or control information.

The dummy burst has the same structure like the normal burst with the difference that no useful data is

transmitted. The dummy burst is necessary because the MS monitors the signal strength of

neighbouring BSs during a call in order to get information for handovers. For this reason every BS has

to transmit all the time on the broadcast channel with full power.

The transmitted power during a burst has to fit in a time mask in order not to interfere data transmitted

from other MS using neighbouring time slots. In Figure 1-6 you can see the time mask for the normal


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duration bursts how it is specified by ETSI. The time mask for the access burst is shorter because

when the mobile sends this burst the timing advance is not adjusted.

Figure 1-6: Time mask for the normal duration bursts [Dav96]

1.4.2 Logical channelsIn order to work properly a mobile radio system has to transmit several informations over the radio

channel. Because of its specific functions this information can be classed into several logical channels.

There is a principal differentiation between Traffic Channels (TCH) and Control Channels (CCH).

Further the CCH is divided into Broadcast Control Channel (BCCH), Common Control Channel

(CCCH) and Dedicated Control Channel (DCCH). The DCCH serves similar tasks like the ISDN

D-Channel, whereas the other two channels have mobile radio specific tasks. An overview over all

traffic and control channels is given in Table 1-1.

Logical Channels

TCH(Traffic Channel, duplex) CCH(Control Channel)

FEC1-coded

Speech

FEC-coded

Data

BCCH

Broadcast

CCH

CCCH

Common

CCH

DCCH

Dedicated CCH

BSMS BSMS BSMS SDCCHStand-Alone

DCCH

BSMS

ACCH

Associated

CCH

BSMSTCH/F

22,8 kbit/s

TCH/F9,6

TCH/F4,8

TCH/F2,4

22,8 kbit/s

FCCH

Frequency

Correction

Channel

PCH

Paging

Channel

BSMS

SDCCH/4 Fast ACCH

FACCH/F

FACCH/H

TCH/H

11,4 kbit/s

TCH/H4,8

TCH/H2,4

11,4 kbit/s

SCH

Synchron.

Channel

RACH

Random

Access Ch.

MSBS

SDCCH/8 Slow ACCH

SACCH/TF

SACCH/TH

SACCH/C4

SACCH/C8

AGCH

Access Grant

Channel

BSMSTable 1-1: Overview Traffic- (TCH) and Control Channels (CCH) [Dav96]

1Forward Error Correction


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The TCHcarries digitally encoded speech or data. Full and half rate TCHs are specified in GSM. The

different data rates for data transmission (9,6 kbit/s, 4,8 kbit/s and 2,4 kbit/s) are achieved by using

different coding algorithms for error detection and correction.

The BCCHis unidirectional from the BS to the MS and supplies the MS with following data neededfor the communication with the BS:

Configuration of the Common Control Channel Information about the frequency mapping at the BS Information about the location of the BCCH in neighbour cells Optional information about Frequency Hopping (FH), Voice Activity Detection (VAD) and power

control

Radio criterions for the cell selection, e.g. minimum received field strength.The BCCH is organised in a multiframe consisting of 51 frames (see Figure 1-5) and is transmitted in

the zeroth time slot of a carrier without frequency hopping and without power control. The reason why

for this carrier no power control is used is that MS located in other cells listen to the BCCH and the

strength of its carrier is a measure of the path loss which is needed for handovers.

Further the BCCH carries the Frequency Control Channel (FCCH) for frequency correction using the

frequency correction burst and the Synchronisation Channel (SCH) for synchronisation using the

synchronisation burst [Dav96].The CCCH is used for setting up calls. The MS initiates a call by sending an access burst on theRandom Access Channel (RACH). If there are free resources the MS is informed on the Access Grant

Channel (AGCH) which Traffic Channel (TCH) and Slow Dedicated Control Channel (SDCCH) to

use.

Is there a incoming call for a MS the BS sends this information on the Paging Channel (PCH).

The DCCHserves similar functions like the ISDN D-Channel and some mobile radio specific taskslike transmitting of measurement data. It is divided into the Stand Alone DCCH (SDCCH) and the

Associated Control Channel (ACCH).

The SDCCH is always used when there is no TCH assigned to the MS. His tasks are

Informing the MS which channel to use Transmitting of billing data Location updating and call forwarding Call set up

The ACCHis used when a TCH is assigned. It is divided into the Fast ACCH (FACCH) and the SlowACCH (SACCH). The FACCH is used when control information has to be transmitted at a high rate

(e.g. during a handover). For transmitting the FACCH the TCH is used which is marked by setting the

stealing bits. The SACCH is used for exchanging control information at a low rate (e.g. power control,

timing advance and quality measures).

1.4.3 Frequency HoppingAn option for GSM network operators is to implement slow Frequency Hopping. In contrast to fast

frequency hopping as it is used for military proposals where the hopping frequency is higher than the

bit rate allows the specified frequency hopping in GSM only a change of the frequency after eachburst.

There are two benefits using frequency hopping. First there is frequency diversity. As discussed in

Section 1.6 is the transmitted information spread over several bursts and even if one burst has a very

high bit error rate due to a deep fade is it possible to determine the correct data bits due to the

information in the other bursts. This can be used for diversity. Since the chosen frequencies have to be

uncorrelated is the probability very high that a slow moving user who is in a deep fade during one

burst using one frequency will not be in a deep fade in the following burst where another frequency is

used. In order that the fast fading faced by two frequencies is uncorrelated the frequencies must be

separated at least by the coherence bandwidth. The coherence bandwidth is defined as the maximum

frequency difference for which two signals have a certain value of correlation [Par92]. The coherence

bandwidth in an indoor environment for a correlation coefficient of 0,5 is approximately 5MHz which

limits the possibility of using frequency hopping in GSM as frequency diversity scheme as the whole

downlink band in EGSM is just 25MHz wide.


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The second benefit of frequency hopping is interferer diversity. Here is a frequency channel with a

very weak C/I-ratio shared by many calls which use the weak channel cyclical. So the mean C/I-ratio

for all calls is lower but this is for capacity reasons better than having a lower number of calls with a

very good C/I-ratio whereas a whole carrier cannot be used through its bad C/I-ratio.

Figure 1-7 shows the algorithm used for determining the hopping sequence in GSM. There are several

input variables for this algorithm. First there is MA the set of RF-channels called mobile allocation.

The MA contains N radio frequencies with 1N64. The mobile allocation index offset(1MAIO1). A Further input is the frame number (FN) in terms of T1, T2, T3 which aredetermined using equations 1.1 to 1.3 (mod stands for the modulo operation).

64modFN1T = (1.1)

26modFN2T = (1.2)

51modFN3T = (1.3)

The Hopping Sequence Number (0HSN64) specifies the hopping sequence to use. All this

information is broadcast over the BCCH and the SCH.The function RNTable simply assigns one out of 114 pseudo-random numbers specified by GSM

according to its argument. NB stands for the number of bits which are necessary to express the number

of RF-channels N. The XOR operator means bit wise exclusive OR, while the remaining functions are

self explaining.

M=T2+RNTable[(HSN XOR T1)+T3]

M'=M mod 2NB

T'=T3 mod 2NB

HSN=0?

No

Yes

M'


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1.5 Frame Structure 8

1.4.4 Discontinuous TransmissionIn order to reduce interference and to lengthen the battery life-time of handsets Discontinuous

Transmission (DTX) is specified as an option for up and downlink in GSM. This means that the

network operator can decide if they want to use DTX or not. If DTX is activated transmits the

transmitter in every burst only if the user is speaking. Otherwise is the data transmission reduced to 1

burst every 480ms. These 2 bursts per second describe the background noise. The background noisehas to be transmitted in order that the listener at the other end of the line does not feel like having an

interrupted connection. Therefore the noise information is fed into a noise generator which creates the

so called comfort noise.

1.4.5 Power ControlAnother possibility to increase the battery life-time of the handsets and to reduce interference on the

radio channel is to control the transmitting power of the base station and the mobile station. Like

frequency hopping and discontinuous transmission is it up to the network operator to decide whether

power control is implemented on the up or downlink or in both directions.

If power control for the downlink is used, the BTS should be able to reduce its transmitting power

down to 30dB of the maximum transmitting power in 15 2dB steps. Also the MS should be able tocontrol the RF-power between its maximum and minimum in 2dB steps.

The decision with which power to send is made in the base station system according to quality

measures made in the BS and MS. The transmitting power must not change more than 2dB every

60ms. Therefore if e.g. a change from 17dBm to 37dBm is requested this will need 600ms. Figure 1-8

shows the adjusting of the mobile's transmitting power through commands of the base station system.

The initial power levels to use in the MS are transmitted in the BCCH.

transmission level(dBm)

39

13 time(60ms intervals)

commands: 17dBm 37dBm 35dBm

Figure 1-8: Transmission power adaptation

1.5 Frame StructureThe in Section 1.4.2 defined logical channels are transmitted over one physical channel. Hereby the

logical channels are arranged in a frame structure where the number of bursts during a frame used byone logical channel depends on the data rate of the logical channel.

The frame hierarchy in GSM knows 4 types of frames (shown in Figure 1-9):

TDMA frame Multiframe Superframe Hyperframe

The TDMA frame consists of 8 time slots, which are dedicated to one channel. 26 or 51 TDMA

frames form one multiframe. Note that the 25thframe of the 26 multiframe is not used. This idle frame

can be used for measuring noise and interfering signals.

A superframe is built of 51 26 multiframesor 26 51 multiframesand 2048 superframes build onehyperframe with a duration of 3h 28min 52s 716ms.


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1.6 Channel Coding 9

Figure 1-9: The hierarchical structure and duration of the different frames in GSM

1.6 Channel CodingThe term channel coding means the adapting of the data bits to be transmitted to the transmission

channel. This contains actions for error correction and modulation.

Actions for error correction are coding and interleaving.

1.6.1 CodingIn GSM there are 3 different codes:

Convolutional codes are used for error correction purposes. The specified convolutional code hasa length of 5 and code rate .

Fire codes are used to detect bursty errors. These are errors which occur in groups. The fire codeused in GSM is able to correct 11 consecutive faulty bits [Mou92].

The parity code is a simple block code which is used for error detection [Meh97].

1.6.2 InterleavingIt is necessary to spread related data blocks over several bursts because error correcting codes are

better in detecting single bit errors whereas in a mobile communication environment mainly burst

errors due to fading occur. In GSM speech data is spread over 8 bursts and data traffic channels arespread over up to 19 bursts.

Figure 1-10 shows the interleaving for a speech channel. The 456 coded data bits of a block are

written row by row in a 8 column by 57 row matrix. In this way 8 subblocks with 57 bits each are

created. In Figure 1-10 there are shown 3 consecutive data blocks (A, B and C) whose bits are grouped

into 8 subblocks with 57 bits each (marked in block B). The numbering scheme for the subblocks can

be seen at data block A. Note that consecutive bits in the original blocks are in different subblocks.

The subblocks are spread over 8 consecutive bursts using a technique called diagonal interleaving.

This results in bursts containing 2 subblocks of different data blocks. The bits of the subblocks 0 to 3

use even bit positions in the data bursts and the bits of the subblocks 4 to 7 use the odd bit positions.

So the 4 first bursts share data block B with the previous data block (block A) and the last 4 bursts are

shared with the consecutive data block (block C).


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11

Chapter 2

Radio Wave Propagation

This chapter gives a brief theoretical description of radio wave propagation where the main interest is

on the physical behaviour of the signals in the mobile channel. A few effects are not described very

deeply as they have only a slight influence to the problem investigated in this project.

The term radio wave is used generic in this chapter and means a TEM-wave in the far-field as far as

nothing else appears from the context. I.e. the distance to the emitting antenna is further than

=

2D2

R (2.1)

where D and are the largest dimension of the antenna and the wave length, respectively. R is knownas Rayleigh distance in the literature [Bon97].

There are many modes of propagation which mainly depend on the used frequency (E.g: ionospheric-,

tropospheric- or ground waves).

Frequencies in the range from 880MHz to 2GHz are interesting for this project as this frequency band

covers the frequency bands of GSM900, DCS1800 and PCS1900. This frequency band belongs to the

UHF band which covers the frequency range from 300MHz to 3GHz [Par92]. UHF waves propagate

normally as ground waves [Par92].

2.1 The Physics of Radio Wave PropagationThis section deals with the physics of radio propagation which is quite similar to the behaviour of

light. This is not very surprising as the only difference between radio waves and light is the higher

frequency of the light.

First the concepts of reflection, diffraction and scattering are discussed and then the free space

propagation will be discussed.

2.1.1 Reflection

If a radio wave which propagates in one medium impinges on another medium having differentproperties then a part of the wave will be reflected and another part will continue the propagation in

the other medium. Is the second medium a perfect dielectric (conductivity =0) than there will be nolosses through absorption in the second medium. Is the second medium a perfect conductor

(conductivity =) than the entire impinging wave will be reflected to the first medium.


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2.1 The Physics of Radio Wave Propagation 12

Figure 2-1: Geometry for calculating the reflection coefficients between two dielectrics. The subscripts i, r, t,

refer to the incident, reflected, and transmitted fields. Parameters 1, 1, 1and 2, 2, 2represent the

permittivity, permeability and conductance of the two media.

Figure 2-1shows a radio wave impinging on the boundary between medium 1 and 2. A part of the

energy is reflected into medium 1 with the angle

ir = . (2.2)

Another part is refracted with the angle tto medium 2. The angle tis given by Snell's law [Rap96]

( )][

90sinarcsin90

22

i11t

= (2.3)

where 1, 1, 2and 2are the permeability and permittivity of the two media, respectively. The fieldstrength of the transmitted and reflected wave (Etand Er) are given by [Rap96]

ir EE = (2.4)

( ) it E1E += (2.5)

where is the reflection factor or || according to the orientation of the E-field. The reflectioncoefficient is a function of the angle of incident, the polarisation of the impinging wave and the

properties of the two media. The reflection factor for the two cases of parallel and perpendicular

E-field polarisation at the boundary of two dielectrics are given by [Rap96]

( ) ( )

( ) ( )t1

1i

2

2

i

1

1t

2

2

||

sinsin

sinsin

+

= (E-field in plane of incidence) (2.6)

( ) ( )

( ) ( )t11

i2

2

t

1

1i

2

2

sinsin

sinsin

+

= (E-field not in plane of incidence) (2.7)


26/121


2.1.2 DiffractionBecause of diffraction is it possible that radio waves propagate along the curved surface of the earth

beyond the horizon or in shadowed areas. This phenomena can be explained with the help of Huygen's

principle which says that every point of a wave front is the source of a secondary spherical wave. The

field strength in a shadowed area is the vector sum of all secondary waves. Figure 2-2 shows this

phenomena.

Figure 2-2:Principle of diffraction [Bla93]

2.1.3 ScatteringIf a plane radio wave impinges on a rough surface then is it spread out in all directions. In this case is

equation 2.2 still valid. Since the surface has a lots of different orientations is the incident wave

reflected in different directions (see Figure 2-3).

Figure 2-3:Principle of scattering [Bla93]

If a surface is rough or smooth can be tested with the help of the Rayleigh criterion, where the critical

height of the surface protuberances (hc) is given by [Rap96]

( )ic

sin8h

= . (2.8)

A surface is called smooth if its minimum to maximum protuberance are smaller than h c. Otherwise

the surface is called rough. The reflection factor for smooth surfaces has to be multiplied by anscattering loss factor if the surface is rough [Rap96].


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2.1.4 Free Space PropagationThe easiest mathematical describable propagation form is the free space propagation. In this case is

assumed that the transmitting antenna is placed in free space. It is assumed that the antenna has a gain

GTin the direction of the receiving antenna which is also placed in free space. The power density per

unit area in a point with the distance d is then

2

TT

d4

GPW

= (2.9)

where PTrepresents the transmitting power. If the receiving antenna has an effective area A then is the

received power

=

=

4

G

d4

GPA

d4

GPP R

2

2

TT

2

TTR

(2.10)

where GRrepresents the gain of the receiving antenna in the direction of the transmitting antenna. The

relationship between transmitted and received power is given by

2

RT

T

R

d4GG

P

P

= (2.11)

This is a fundamental relationship which is known in literature as Friis equation [Par92]. The free

space propagation loss obeys an quadratic square law with range d.

Figure 2-4: Propagation over a plane earth (T and R stand for transmitting and receiving antenna)

Figure 2-4shows the configuration when the antennas are mounted in a height hTand hR(superscript T

and R stand for transmitting and receiving antenna) over plane earth. Assuming that the earth is an

ideal conductor and that d>>hT, hRfollows the relationship between transmitted and received power as

[Par92]

2

2

RTRT

T

R

d

hhGG

P

P

= (2.12)

This a equation shows an inverse fourth-power law with the range. This is very close to what can be

measured in real environment where the path loss PL can be expressed as a function of distance by the

power law using a path loss exponent [Rap96]

0d

dPL (2.13)

where d0 is the close-in reference distance which is determined from measurements close to the

transmitter and d is the separation between receiver and transmitter. In urban environments is the path

loss exponent typically between 3 and 5.


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2.2 Multipath Propagation 15

The real mobile environment is too complicated to calculate the path loss deterministic. Therefore is

the path loss described by models which are derived by analytical and empirical methods. The

empirical approach is based on fitting curves or analytical expressions that recreate a set of measured

data. This has the advantage of implicitly taking into account all propagation factors, both known and

unknown, through actual field measurements. Several models have established and are used to predict

large-scale coverage for mobile communication systems design (e.g. Egli model, JRC method,

Blomquist-Ladell model to name some). A detailed description of these models is beyond the scope of

this report. The interested reader is referenced to [Par92].

2.2 Multipath PropagationIf a radio wave propagates through the mobile environment then the signal received at a receiving

antenna is composed of components which origin to different phenomena like diffraction, reflection

and refraction. The term for this is multipath fading.

Figure 2-5 shows the received signal strength which was measured in an indoor environment with a

handset moving on a predefined path. The path losses can be split up into two parts. First there are the

losses according to shadowing which is called slow fading. On the other hand the signal strength

varies because the incoming waves have travelled over different distances and therefore they have

different phases. These signal components can interfere in a constructive or destructive way which is

called fast fading.

0 1 2 3 4 5-12

-10

-8

-6

-4

-2

0

2

4

6

Time [s]

ReceivedP

ower[dBovermean]

measured signal strengthslow fading

Figure 2-5: Received signal strength and slow fading in an indoor environment

2.2.1 Slow FadingSlow fading is often called shadowing because hills and buildings are shadowing the radio wave. As

the mean path loss is log-normal distributed another often used term for this kind of fading is log-

normal fading. The dashed line in Figure 2-5 shows the trend of the slow fading in an indoor

environment.

The received signal can be expressed as

)t(r)t(m)t(E o= (2.14)

where m(t) and ro(t) represent the slow and fast fading, respectively.

The slow fading is extracted of the measured signal strength by building the local mean over a certain

length. The length of the averaging window has to be adjusted to the environment. If the windowlength is chosen to short then the slow fading will still contain some parts of the fast fading. Otherwise

if the window length is to long then the calculated fast fading will still contain parts of the slow fading,


29/121


which will change the probability distribution of the signal. E.g. a Rayleigh distribution will be

changed into another distribution.

2.2.2 Fast FadingFast fading is also known as short-term fading because due to the fast fading the signal strength at an

antenna can change dramatically when the antenna is moved over a very short distance. The fastfading is caused by signal components coming to the receiving antenna from different directions. As

they have travelled over different distances the phase of the signals at a certain receiving point will be

randomly distributed. These signals will interfere constructively or destructively according to their

phase relations. A typical distance between constructive and destructive interference is /2 where isthe wavelength of the radio wave.

The fast fading rois calculated using equation 2.15

)t(m)t(r)t(ro = [dB] (2.15)

where r(t) and m(t) are the measured signal and the slow fading, respectively. Figure 2-6 shows the

fast fading in an indoor environment.

0 1 2 3 4 5-8

-6

-4

-2

0

2

4

6

Time [s]

astFading[dB]

Figure 2-6: Fast fading in an indoor environment

Figure 2-7: The co-ordinate system of the scattering model


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The distribution of amplitude and phase of the received signal can be deviated using the scattering

model. Figure 2-7 shows the co-ordinates of the scattering model. In this model it is assumed that the

received signal is composed of a big number of components with amplitude Cn, phase nand spatialangles nand nwhich are random and statistical independent. The quadratic mean amplitude is givenby

[ ]N

ECE 0

2n = (2.16)

where E0 is a positive constant and N represents the number of incoming waves. The received field

strength E(t) is

=

=N

1n

n )t(E)t(E (2.17)

with

( ) ( ) ( ) ( ) ( )[ ]nn0nn0nn0onn sinzcossinycoscosx2

tcosC)t(E +++

= (2.18)

where x0, y0 and z0 represent the position of the receiving antenna in the co-ordinate system and

0=2f0, where f0is the frequency of the radio waves.If we assume that the receiver is moving with the speed v in the xy-plane in a direction enclosing the

angle with the x-axis then the co-ordinates of the receiver are given by

( )

( )

.constz

sinvy

cosvx

0

0

0

=

=

=

(2.19)

The received field strength can be written as

( ) ( )tsin)t(Qtcos)t(I)t(E 00 += (2.20)

where I(t) and Q(t) are the in-phase and quadrature components that could be received by a suitable

receiver.

( )=

+=N

1n

nnn tcosC)t(I (2.21)

( )=

+=N

1n

nnn tsinC)t(Q (2.22)

with

( ) ( )nnn coscosv2

= (2.23)

( ) nn0

n sinz2

+

= . (2.24)

n=2fnis the Doppler shift experienced by the n thcomponent. If z0is different from 0 then the firstpart of equation 2.24 is the projection of the phase into the phase reference lying in the xy-plane.

If there is a high number of incoming waves and there is no dominating wave then follows by thecentral limit theorem that I(t) and Q(t) are independent Gaussian processes. As the mean values of I(t)

and Q(t) are zero, follows that the mean value of the envelope is also zero. I(t) and Q(t) have the same


31/121


variance 2which is equal to their mean power. The PDF of the in-phase and quadrature componentcan be written as

2

2

2

x

x e2

1)x(p

= (2.25)

where x = I(t) or Q(t) and 2=E0/N.The envelope r(t) and the phase (t) are given by equations 2.26 and 2.27.

)t(Q)t(I)t(r22 += (2.26)

=

)t(I

)t(Qarctan)t( (2.27)

Since like mentioned before I(t) and Q(t) have zero mean and the same variance is the joint probability

density function pIQ:

QIIQ ppp = (2.28)

2

22

2

QI

2IQe

2

1)Q,I(p

+

= (2.29)

Applying a co-ordinate transformation from pIQ(I,Q) to pr(r,) we get the joint PDF pr(r,)

22

1

2re

2

r),r(p

= . (2.30)

The PDF of the phase pis derived by integrating pr(r,) over the envelope r.

==0

r

otherwise0

202

1

dr),r(p)(p (2.31)

As you can see is the incoming phase uniform distributed. In the same way we get the PDF of the

envelope pr.

2

2r2

02rre

rd),r(p)r(p

==(2.32)

Equation 2.32 is well known as Rayleigh distribution. The mean value E[r], the mean square value

E[r] and the standard deviation Aof the Rayleigh distribution are given by

[ ]

==

0

r2

dr)r(rprE (2.33)

[ ] 20

r22

2dr)r(prrE ==

(2.34)

[ ] [ ] ==2

2rErE22

A . (2.35)


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Figure 2-8shows the PDF of the Rayleigh distribution.

Figure 2-8: PDF of the Rayleigh distribution; 1=median (50% value), 1,1774, 2=mean value, 1,2533,

3=RMS value, 1,41

Rician FadingIn the deviation above we assumed that there is no dominating signal like it is in a non line of sightsituation. Is there a line of sight between transmitting and receiving antenna then there will be one

dominating signal. Therefore the mean I(t) and Q(t) will be different from zero and there will be less

deep fades. In this case the joint PDF of pr(r,) is according to [Par92]

( )2

s2s

2

2

cosrr2rr

2re

2

r),r(p

+

= (2.36)

where rsis the envelope of the dominant signal. By integrating over we get the PDF of the envelopepr(r).

= +

2

s0

2

rr

2r

rrIe

r)r(p

2

2s

2

(2.37)

I0(.) is the modified Bessel function of the first kind and zero order, which is given by

=

=0n

n2

n2

0!n!n2

x)x(I . (2.38)

The distribution function defined in equation 2.37 is called Rician distribution. Therefore this kind of

fading is often referred to as Rician fading.An alternative form of the Rician distribution is given by equation 2.40, where the Rician factor K

(equation 2.39) represents the ratio of the power in the dominant signal to the power in the multipath

(random) components.

=

2

2s

2

rlog10K (2.39)

( )

=+

2s

10

K

0

rrr

10

2s

10

K

rr

10r2Ie

r

10r2)r(p

2s

2

2s

10

K

(2.40)


33/121


Figure 2-9shows the PDF of the Rician distribution for different values of K. If K goes to 0 the PDF

becomes the form of the Rayleigh distribution and if K>>1 then the Rician distribution looks like a

Gaussian distribution with mean rs.

Figure 2-9: PDF of Rician distribution; (a) K

0, (b) K 1, (c) K >> 1

The PDF of the phase p() at the presence of a dominating signal results by integrating equation 2.36over the envelope r [Par92].

( ) ( ) ( )

+

+

=

2

cosrerf1e

cosr

21e

2

1)(p s2

cosr

s2

r2

22s

2

2s

(2.41)

erf(.) stands for the error function which is given by

=x

0

tdte

2)x(erf

2

. (2.42)

If rs/ tends to zero then the resulting phase will be uniform distributed in the interval [0;2[. If

rs/>>1 then the phase will be determined by the phase of the dominating signal.

2.2.3 Doppler ShiftIf the distance between transmitter and receiver changes then the phase position of the received signal

will also change. Figure 2-10 shows a mobile receiver which is moving with the speed v from A to B.

During the time t travels the receiver a distance d=vt. This leads to a change of the path lengthbetween transmitter and receiver of l=dcos(). The received phase changes in the time t by

( )

= costv2

. (2.43)

This means that the phase changes over the time, which can be expressed by a frequency shift. Thisfrequency shift fDis called Doppler frequency and can be expressed by

( )=

= cosf

dt

d

2

1f mD (2.44)

where fmis the maximum Doppler frequency given by

=

vfm (2.45)


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Figure 2-10: Illustration Doppler effect [Dav96]

The extreme values of the Doppler frequency (fD=+fm and fD=-fm) result when the mobile station is

moving directly towards the transmitting antenna or in the opposite direction.

In the case that the moving mobile station receives a lot of different signal components from different

directions then the signal components will experience a different frequency shift according to their

direction . Because of this a continuous wave transmitted from a base station will have a spreadedspectrum of the bandwidth 2fmat the moving receiver. This spectrum has a specific form according to

the environment and the antenna characteristics and is called Doppler spectrum.

Assuming that the received signal is composed of many components (like in the chapter before) so that

the power density is continuous distributed in the area [,+ d] then is the power coming from thisdirection p(). Using a receiving antenna with a horizontal directivity pattern G2() results in areceived power S()in the angle area d

= d)(p)(AGd)(S2

(2.46)

where A is a constant which depends on the path loss and the transmitting power of the base station.

The power spectrum

2

m

0m

2

f

ff1f

)(G)(Ap)f(S

=

(2.47)

results by a variable transformation. f0 in equation 2.47 is the transmitting frequency. Assuming the

incoming components are uniformly distributed over the angle area =[0; 2[ and a vertical monopolewith an omnidirecitonal horizontal directivity pattern G

2()=1.5 as receiving antenna the Doppler

spectrum is given by equation 2.48 and will have the bath tube form shown in Figure 2-11.

2

m

0m

f

ff1f4

A3)f(S

=(2.48)

The deviation above is assumed on the Clark model, where uniformly distributed incoming signals

over the angle area =[0; 2[ with an elevation angle =0 (horizontally waves) are assumed. Models

with more complicated distributions of the incoming signals can be found in [Par92] page 116 to 120.


35/121


Figure 2-11: Doppler spectrum of a non-modulated carrier

2.2.4 Delay SpreadBecause of the multipath propagation will the mobile station receive signals with different delays

which means that there is time dispersion. A short impulse (t) transmitted from a base station willlead to a certain number of impulses with different attenuation at the mobile station. The channel

impulse response h(t) can be written as

=

=N

1n

nn )t(a)t(h (2.49)

where anand n is the attenuation and the delay of the n thsignal. N is the total number of incomingsignals. Typical channel impulse responses for different areas are shown in Figure 2-12. If the nhaveabout the length of the bit duration there will be inter symbol interference which makes it more

difficult to detect the transmitted information. Equalisers are used to reduce this problem. For example

the equaliser in a GSM handset must be able to deal with delays of up to 16 s or 4 bits.

A measure for the in a channel impulse response occurring delays is the delay spread which isdefined as the second central moment of the delay power spectrum |h(t)|

2:

( )

=

dt)t(h

dt)t(htt

2

22

(2.50)

with the mean access delay

=

dt)t(h

dt)t(htt

2

22

. (2.51)


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25

Chapter 3

Diversity and Combining Methods

The term diversity means the quality of having variety. In the context of radio signals this means

having choice between different signals. The basic idea is to get the transmitted information from

different statistical independent fading channels. The probability that there is a deep fade at the same

time in two uncorrelated propagation paths is very low. This can be seen in Figure 3-1.

0 0.5 1 1.5 2 2.5-12

-10

-8

-6

-4

-2

0

Time [s]

ReceivedPower[dBbelowmaximum]

Patch AntennaDipol Antenna

Figure 3-1: Received signal strength of patch and dipol antenna in an indoor environment (correlation

coefficient of the fast fading FF=-0,2)

Some diversity schemes are mentioned in the following section.

3.1 Different Diversity Schemes

3.1.1 Space DiversitySpace Diversity can be used in base stations and in mobile equipment. The idea is that the fast fading

at two separated antennas is uncorrelated if the antennas have a certain distance. The necessary

distance between the antennas is significantly larger for base stations due to the different surroundings

of mobile and base station. Normally the antenna of a base station is mounted much higher than the

height of the mobile antenna. Therefore the base station antenna is clear of its surroundings whereas

the mobile unit antenna is embedded in them. Figure 3-2 show the different environments for base and

mobile station.


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3.1 Different Diversity Schemes 26

Figure 3-2: The different environments at the base station and the mobile station

The correlation of the signal envelope r(d) between two separated antennas is [Lee98]

=

d2J)d(

20r (3.1)

where d is the distance between the antennas, is the wavelength of the radio wave and J0(.) is theBessel function of the first kind zeroth order. In Figure 3-3 r(d) is plotted over d/. The firstminimum of r(d) is at d0,4. This means that two identical antennas with the same polarisation of amobile operating in the 1800 MHz band should be separated by 6 cm.

0 0.5 1 1.5 20

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

1

Normalisedcorrelationcoeff

icient

d/Figure 3-3: The normalised correlation coefficient over the separation of the antennas

3.1.2 Polarisation DiversitySignals transmitted in different polarisations will exhibit uncorrelated fading statistics in a mobile

radio environment. This can be used in a diversity scheme. Therefor the mobile unit needs two

antennas which are orthogonally polarised. Using polarisation diversity does the distance between the

two antennas not matter. In non line of sight conditions it is not necessary to transmit in both

polarisation because there will exist both possible polarisations at the mobile station through the cross

coupling in the mobile radio environment.


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3.1 Different Diversity Schemes 27

3.1.3 Pattern DiversityAnother possibility to get uncorrelated signals at the mobile station is to use antennas with different

antenna patterns. Figure 3-4 shows the antenna patterns in the XY-plane for the dipole and patch

antenna of the handset used for measuring the channel data for this project. It can be seen that the

antenna patterns for both polarisations are quite different.The correlation of the fast fading will be low because the incoming signals will be weighted in a

different way according to their angel of arrival and polarisation.

a)

5

10

15

20

25

30

35

30

210

60

240

90

270

120

300

150

330

180 0

30 correspond to 0dB

b)

5

10

15

20

25

30

35

30

210

60

240

90

270

120

300

150

330

180 0

30 correspond to 0dB

Figure 3-4: XY-plane of the simulated antenna patterns of the dipole (a) and patch (b) antenna of the modified

test handset; dashed line represents polarisation E, solid line represents polarisation Eand dotted line

represents |E|+|E|; (user present looking towards 270, handset inclined by 60)

3.1.4 Frequency DiversityWhen frequency diversity is used the same information is transmitted over several carriers with

different frequencies. If the used radio environment is frequency selective then the sufficiently spaced

carriers will face uncorrelated fading. The correlation of the transmission coefficient for two

frequencies is statistical dependent on their distance. This dependency can be evaluated by applying

the Fourier transformation to the auto correlation of the mean impulse response [Par92]. Figure 3-5

shows the frequency correlation over frequency separation in an indoor environment. In this case the

frequency separation should be at least 4,6 MHz if a correlation of 0,5 is assumed to be less enough to

achieve a sufficient diversity gain.

One benefit of frequency hopping which is used by some GSM network operators is frequency

diversity. As discussed in Section 1.6.2 is the transmitted information spread over several bursts and

even if one burst has a very high bit error rate due to a deep fade is it possible to determine the correct

data bits due to the information in the other bursts. Since the chosen frequencies have to be

uncorrelated is the probability very high that a slow moving user who is in a deep fade during one

burst using one frequency will not be in a deep fade in the following burst where another frequency is

used.


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3.2 Combining Techniques 28

0 5 10 15 200

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

1

Frequency separation [MHz]

Frequencycorrelation

Figure 3-5: Frequency correlation over frequency separation in an indoor environment

3.1.5 Time DiversityIn a system using time diversity the same information is transmitted at different times. The channel is

uncorrelated for two different time points when the mobile station is moving because the distance

between two fades is very short. The performance of time diversity increases with increasing speed of

the mobile unit. Note that no additional hardware is necessary for using time diversity.

3.2 Combining TechniquesThe gain achieved by using a diversity scheme depends a lot on the method used for combining the

received signals. For the discussion of the different combining techniques we assume space,

polarisation or pattern diversity where a several number of antennas is used.

3.2.1 Switched CombiningSwitched Combining is the combining technique with the lowest effort in hardware. Only an antenna

switch is needed. The algorithm switches to another antenna if the signal quality falls below a certain

threshold. The threshold is determined as a kind of local mean.

Antennas

Switch

ReceiverRX

Antennas

Switch

Receiver

Output

Figure 3-6: Switched Combining


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3.2 Combining Techniques 29

3.2.2 Selection CombiningA receiver for each antenna is necessary if selection combining is used. The signal with the highest

signal to noise ratio (SNR) is selected in the base band for further use. The SNR of selection

combining is never higher than the SNR of the best signal because the signals of the non-chosen

antennas are discarded.

Antennas

ReceiverRXRX RX RX

Selection Logic

Output

...

Figure 3-7: Selection Combining

3.2.3 Maximum Ratio CombiningA system using maximum ratio combining weights the signals of the antennas according to their SNR,

aligns their phases in the base band and adds them. This method is the best combining technique if

only noise is present. A drawback of maximum ratio combining is the huge amount of calculations

which are necessary to determine the correct weight setting.

Antennas

Adjustable Amplifiers

RXRX RX RX

Output

...

phase correctsummation and

weight generation

Receiver

Figure 3-8: Selection Combining

3.2.4 Equal Gain CombiningA simpler method than maximum ratio combining with nearly the same performance is equal gain

combining. This combining method aligns only the phases of the base band signals before addingthem.

3.2.5 Optimum CombiningAll the combining techniques mentioned above improve only the SNR. Optimum combining has the

same structure like maximum ratio combining. The difference is that in optimum combining the

weights are chosen in that way that the signal to interferer and noise ratio (SINR) is maximised.


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3.3 Model For Optimum Combining 30

3.3 Model For Optimum CombiningIn this paragraph the principle of optimum combining is explained and a mathematical relation for the

weights is derived.

In Figure 3-9 a block diagram of an M antenna element adaptive array can be seen. The complex

baseband signal received by the i thantenna element in the kthsampling interval xi(k) is multiplied by acontrollable complex weight wi. The array output y(k) is then formed by summing the weighted

signals.

The optimum weights are the ones that minimise the SINR (Signal to Interferer and Noise Ratio).

WeightGeneration

r(k)

y(k) ArrayOutput

ReferenceSignal

(k)- +

wM(k)

w1(k)

x1(k)

xM(k)

...

...

Figure 3-9: Block diagram of an M-element adaptive array

Let the complex weight vector wbe given by

=

M

2

1

w

w

w

w (3.2)

and the received complex signal vector xis given by:

=

M

2

1

x

x

x

x (3.3)

The received signal consists of the desired signal, thermal noise and interfering signals. This can be

expressed as:

=

++=L

1j

jnd xxxx (3.4)

where xd, xnand x

are the received desired signal, noise, and jthinterfering signal vectors, respectively,

and L is the number of interferers. Furthermore, let sd(k) and s

(k) be the desired and jth interfering

signals as they are transmitted, respectively, with

[ ] 1sE 2d = (3.5)


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[ ] .Lj1for1sE 2j = (3.6)

Then xcan be expressed as

=

++=L

1j

jjndd )k(s)k(s uxux (3.7)

where udand u

are the desired and jthinterfering signal propagation vectors, respectively.

The error signal (k) is given by

)k(y)k(r)k( = (3.8)

where r(k) is the kthsample of the reference signal and y(k) is the output of the antenna array at the k

th

sample. Writing y(k) as the weighted sum of the input signals (3.8) becomes

)k()k(r)k(Txw= (3.9)

where superscript T denotes transpose. The square absolute value of (k) is:

( ) wxxwxw )k()k()k()k(r2)k(r)k(r)k( T*T*T*2 += (3.10)

where the superscripts * denotes conjugate. Taking the expected values of2

)k( gives

[ ] wRwrw xxTxrT*2 2)k(r)k(r)k(E += (3.11)

where the line above r(k)r*(k) means the mean value of this expression andrxris

=

)k(r)k(x

)k(r)k(x

)k(r)k(x

*M

*2

*1

xr

r (3.12)

and Rxxis the received signals (desired and interfering signals plus noise) correlation matrix and whichis given by

++

++=

==

TL

1j

jnd

*L

1j

jndxx E xxxxxxR (3.13)

Assuming the noise and interfering signals are uncorrelated, it can be shown that

[ ]=

++=L

1j

Tj

*j

2Td

*dxx E uuIuuR (3.14)

where 2 is the noise power and Iis the identity matrix.


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The weights which minimise2

)k( and maximise the SINR are found by solving

[ ]( ) 0)k(E 2w = . (3.15)

Since

[ ]( ) wRr xxxr2w 2)k(E += (3.16)

follows that the optimum choice for the weights must satisfy

xr1

xxrRw = (3.17)

where superscript 1 denotes the inverse of the matrix. In the optimal case where the cross correlationbetween the desired signal and noise and interfering signal is zero, is

*dxr ur (3.18)

Remembering that multiplying wwith a constant does not affect the SINR at the array output we canwrite the equation for the optimum weights

*d

1xxu= Rw . (3.19)

where is a constant.

[Win84] comes up with a slightly different formula for calculating the optimum weights which results

in another scaling of the weights which does not affect the output SINR of the antenna array [Boc99].Winters suggests the use of a noise and interferer correlation matrix which is given by

[ ]=

+=L

1j

Tj

*j

2nn E uuIR (3.20)

Using this matrix Winters calculates the optimum weights by

*d

1nnu= Rw (3.21)

where

is a constant again.

Two different algorithms to gain the optimum weights are mentioned in this project. The first called

algorithm A is based on equations 3.14 and 3.17 whereas the second called algorithm B is based on the

weight formulas from Winters (equations 3.20 and 3.21).


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3.3.1 Algorithm AAlgorithm A is based on equations 3.14 and 3.17. The optimum weights calculated by algorithm A are

given by

xr1xx rRw = (3.22)

where the estimated received signal correlation matrix is given by

)j()j(K

1 TK

1j

*xx xxR

=

= (3.23)

where K is the number of samples used. The estimated received signal cross correlation vector is given

by

)j(r)j(K1

K

1j

*xr == xr (3.24)

The only available reference signal in GSM is the 26 bit long training sequence. As a GMSK

modulated bit has an impulse response which is at least 3 bit long (see Appendix A) are the first two

bits of the received training sequence distorted by the two preceding bits. Therefore is it not possible

to use the first two bits of the training sequence as part of the reference signal. So the reference signal

is just 24 bits long.

Note that algorithm A does not need any information about the interfering signal in order to calculate

the optimum weights but it is necessary to know the received signals at all antennas in order to

calculate xxR and xrr . This makes two complete receiver chains necessary (see Appendix E). The aim

of this project is to find an algorithm which needs only one receiver. Therefore is it not possible to usealgorithm A for this project.

3.3.2 Algorithm BTo deal with the problem mentioned above the following on the formula of Winters [Win84] based

method is used. The estimated noise and interferer matrix is given by

=

+=L

1j

Tj

*j

2nn uuIR (3.25)

and the weights are calculated by

*d

1nn u= Rw (3.26)

where estimated noise power 2 , the estimated propagation vector of the wanted signal d and theestimated propagation vector of the interfering signal jare determined from the output of the antenna

array as described in Chapter 4 Signal Estimation.

In the case of only two antennas is it possible to scale the weights using equation 3.27 in that way that

one weight equals always one. Doing this demands only one controllable amplifier and one phase

shifter in the receiver.

=

= 21

12 w

w

w

1

w~

1~w . (3.27)

The performance of this algorithm and algorithm A is compared in Appendix F.


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35

Chapter 4

Signal Estimation

In the Section 3.3.2 Algorithm B which describes the Algorithm B is shown that it is necessary to

know the noise power and the propagation vectors of wanted and interfering signal at the input of the

antenna system in order to calculate the optimum weights. As these parameters cannot be measured

exactly they have to be estimated. In the first part of this section it is described, how the signal looks

like and how it is composed for the simulation. Then in Sections 4.2 to 4.6 it is mentioned how the

different parameters of the received signal can be estimated and which estimation error is made by

using these methods.

4.1 Signal CompositionThe signal used in the receiver for estimating the parameters needed for calculating the weights

consists of the wanted burst, the signal from the interfering base station and white gaussian noise. The

amplitude and phase of the wanted and interfering signals will be different because both base stations

can transmit with different powers and the propagation channels are different. The amplitude of the

interfering signal will with a high probability be lower than that of the wanted because the distance to

the interfering base station is larger. The bursts of the wanted and the interfering base station will start

at different times because the base stations in GSM systems are not synchronised.

For evaluating the estimation errors by the suggested algorithms, the signals are modelled as follows.

For the interfering and wanted signal are GMSK modulated GSM bursts with 4 samples per bit with

different training sequences used. Figure 4-1 shows the structure and the phase course of one GSM

Burst. The 2 blocks with the 58 random bits stand for the 57 data bits and the stealing bit of a normal

burst in GSM.

Since the base stations in the GSM system are not synchronised it is unlikely that the training

sequences of the wanted signal and the interferer occur at the same time. To take this into account is

the burst of the interfering signal shifted by 50 Bits (see Figure 4-2). If the training sequences of

wanted and interfering signal would overlap the estimation of the signals would be disturbed by the

same bit sequence every time which leads to the same estimation error especially when the noise level

is very low. In this case it is no use to average over a number of bursts to get a realistic measure for the

estimation errors.The noise is modelled by a random signal which has a Gaussian distributed amplitude and a zero mean

value. The phase of the noise is uniformly distributed between 0 and 2.Figure 4-3 shows how the input signal of the estimators is composed of wanted signal, attenuated

interfering signal and noise.


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4.1 Signal Composition 36

0 50 100 150

-900

-720

-540

-360

-180

0

180

360

Bit Number

Phase[]

26 bit TS58 random bits 58 random bits

3 tail bits 3 tail bits GP

b)

a)

TS Training SequenceGP Guard Period (shortened to 2 bits)

Figure 4-1: a) Structure and b) phase course of one GSM Burst1

26 bit TS58 random bits 58 random bits


58 random bits

26 bit TS58 random bits 58 random bits 58 random bits

50 bit

26 bit TS 58 random bits

TS Training SequenceGP Guard Period (shortened to 2 bits)RB Random Bits

3 tail bits

random bitsRB

Original Burst:

Shifted Burst:

Figure 4-2: Construction of the shifted interfering signal

1The sequence of the 148 transmitted bits (3 tail bits + 57 data bits + 1 steeling bit + 26 bit training sequence +

1 steeling bit + 57 data bits +3 tail bits) has the duration of 150 bits because the impulse response of the GMSK

modulator (which lasts several bits) is 3 bits long in the GSM transceiver (see Appendix A for details). So the

last tail bit (bit number 148) will although cause an output at bit number 149 and 150 where actually no data istransmitted. In order to show this are the first two bits of the 8,25 bit long guard period also plotted in the graph.

The guard period does not influence the quality of the results in this chapter because the estimation algorithmuses only the areas where the training sequences are located.


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4.1 Signal Composition 37

0 50 100 1500

0.5

1

1.5

Bit NumberAmplitude

0 50 100 150-180-90

090

180

Bit NumberPhase[]

Trainingsequence 0

(26 Bit)58 random bits 58 random bits


Wante

dBurst

0 50 100 1500

0.25

0.5

Bit NumberAm

plitude

0 50 100 150-180

-900

90180

Bit NumberPhase[]

Noise

+

+

=

Burst

Structure

0 50 100 1500

0.25

0.5

Amplitude

Bit Number

0 50 100 150-180-90

090

180

Bit NumberPhase[]

3 tail bits3 tail bits GP

InterferingB

urst

0 50 100 1500

0.51

1.5

Bit NumberAmplitude

0 50 100 150-180

-900

90180

Bit NumberPhase[]

EntireBurst

GP Guard Period (shortened to 2 bits)

Trainingsequence 0

(26 Bit)46 random bits58 random bits

12rand.bits

Figure 4-3: Composition of input signal for estimators (C/I=10dB, C/N=20dB)2

(the phase is limited to 180)

2In reality the power of the interfering burst can change after bit number 101 (e. g. when the interfering base

station is not transmitting in this time slot or power control is implemented). In this chapter we are onlyinterested in the parts of the bursts, where the training sequences are placed, so there is no effect on the

estimation error if the power of the interferer is chosen equal for both overlapping time slots. The case ofdifferent interferer powers is mentioned in Section 4.5.2.


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4.2 Propagation Vector of Wanted Signal 38

4.2 Propagation Vector of Wanted SignalThe propagation vector (amplitude and phase) of the wanted signal is calculated by correlating the

received signal with the modulated training sequence from the wanted base station. The phase and

amplitude of the highest correlation peak is taken as an estimate for the propagation vector of thewanted signal.

The received

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GSM Technique